In recent years, the world has witnessed explosive growth in the demand for wireless communications and it is predicted that this demand will increase in the future. In the case of cellular services, for example, there are already over 500 million users that subscribe to cellular telephone services and the this number is continually increasing. Eventually, in the not too distant future, the number of cellular subscribers will exceed the number of fixed line telephone installations. In many cases, the revenues from mobile services already exceeds that for fixed line services even though the amount of traffic generated through mobile phones is much less than in fixed networks.
Other related wireless technologies have experienced growth similar to that of cellular. For example, cordless telephony, two way radio trunking systems, paging (one way and two way), messaging, wireless local area networks (WLANs) and wireless local loops (WLLs). In addition, new broadband communication schemes are rapidly being deployed to provide users with increased bandwidth and faster access to the Internet. Broadband services such as xDSL, short range high speed wireless connections, high rate satellite downlink (and the uplink in some cases) are being offered to users in more and more locations.
In connection with cellular services, the majority of users currently subscribe to digital cellular networks. Almost all new cellular handsets sold to customers are based on digital technology, typically second generation digital technology. Currently, third generation digital networks are being designed and tested which will be able to support data packet networks and much higher data rates. The first generation analog systems comprise the well known protocols AMPS, TACS, etc. The digital systems comprise GSM, TDMA (IS-136) or CDMA (IS-95), for example.
The recent explosion in demand for portable wireless communication devices has prompted a large amount of research in the design of efficient radio frequency (RF) receivers. The requirement of receiver portability, however, limits the available battery power and consequently places severe constraints on the power consumption, physical size and weight of such devices. Therefore, miniature radio receivers that dissipate low power are highly desirable. The search for efficient RF receivers has focused on the development of simplified radio receiver architectures.
For decades, nearly all RF receivers have been based on the well known superheterodyne architecture. In a superheterodyning receiver, an antenna feeds the received RF signal to an RF amplifier. The amplified RF signal is then converted to an intermediate frequency (IF) by mixing it with a signal produced by an offset local oscillator. The resulting IF signal is then amplified by an IF amplifier and then shifted to baseband by mixing it with a signal from a second local oscillator. The baseband signal is then quantized in an analog-to-digital (A/D) converter and demodulated by a digital signal processing (DSP) based demodulator. Alternatively, the radio may use conventional hardware techniques to demodulate the baseband signal and extract the original information signal.
There are several disadvantages of the superheterodyne architecture, however, which make it impractical for low-power implementation. One disadvantage is that in order for an IF amplifier to produce sufficient gain in the IF signal, IF filters biased at large currents are needed which consume substantial power. Furthermore, these IF filters require numerous passive components which cannot be integrated onto a single chip with the rest of the receiver, adding to the size and cost of the receiver. Another significant drawback of the superheterodyne architecture results from the symmetry in mixing the RF signal with the signal from the offset local oscillator. The mixing generates not only the desired RF signal but also undesired image signals at an intermediate frequency above or below the offset local oscillator frequency. Removing the image signals, however, requires a very selective analog RF filter which complicates the design of the receiver and increases its cost.
The direct conversion receiver architecture avoids many of the above difficulties of the superheterodyne architecture. A block diagram illustrating the typical structure of a direct conversion receiver front end circuit is shown in FIG. 1. The receiver, generally referenced 10, comprises an antenna 12 coupled to an analog front end circuit 14 and a DSP based baseband signal processor 16.
The antenna 12 couples the received RF signal to an RF amplifier 18. The amplified RF signal is then converted directly to baseband (hence the term ‘direct conversion’) by mixing it with a signal produced by a local oscillator 22. Both I and Q baseband signals are generated via I and Q mixers 20, 26, respectively. The LO signal without any phase shift is input to the I mixer 20 while the LO signal phase is shifted 90 degrees via phase shifter 24 and is input to the Q mixer 26. The resulting baseband signals are then filtered by low pass filters 28, 32. The filtered analog I and Q signals are then converted to digital by A/D converters 30, 34. The digital I and Q signals are then processed digitally by the baseband processor 16 which demodulates the received signal using a DSP based demodulator.
Note that because the down-converted signal in the direct conversion design is centered at frequency zero, there is no image signal to be rejected making the filtering task relatively easy and thus permitting the use of active low pass filters. In addition, the direct conversion architecture relaxes the selectivity requirements of RF filters and eliminates all IF analog components previously required, allowing for a highly integrated, low-cost and low-power receiver. Due to these and other potential advantages, many vendors have been focusing efforts to designing direct conversion radio receivers.
Several well-known problems associated with direct conversion designs, however, include 1/f noise, multiplicative impairments and additive impairments (also referred to as DC-offset noise). These noise sources result in severe performance degradation of the receiver and reduce the detectability of the transmitted signal.
The 1/f noise (also known as flicker noise or pink noise) is an intrinsic noise phenomenon found in semiconductor devices, with a power spectral density inversely proportional to frequency. The coupling of 1/f noise with the received signal takes place primarily after down-conversion at the baseband amplifier. Since the baseband signal could be in the range of hundreds of microvolts RMS, the 1/f noise can potentially comprise a substantial fraction of the signal power, resulting in large signal distortion.
Note that in a superheterodyne architecture the IF signal is substantially amplified by an IF amplifier. Since the IF frequency is high enough that 1/f noise is negligible, the 1/f noise then becomes relatively insignificant when the IF signal is mixed down to baseband.
The DC-offset noise impairment is an offset voltage that appears in the signal spectrum at DC when an RF signal is converted directly to baseband. This offset value typically may dominate the desired signal and can substantially degrade the signal to noise ratio (SNR) if it is not removed.
The DC offset is derived from three main sources: transistor mismatch in the signal path, the LO signal leaking back to the antenna due to poor reverse isolation through the mixer and RF amplifier and then reflecting off the antenna and self-downconverting to DC via the mixer, a large near channel interferer leaking into the LO port of the mixer and self-downconverting to DC, 1/f noise from any operational amplifiers in the circuit and bias in the filters and amplifiers.
The first source is transistor mismatch in the signal path between the mixer and the I and Q inputs of the detector. With careful circuit design this effect could be largely minimized. The second cause of DC-offset occurs when the signal from the local oscillator, which is at the same frequency as the RF signal, leaks from the antenna and reflects off an external object and self-converts to DC. This local oscillator radiation also interferes with other nearby receivers tuned to the same frequency. Since this radiation may be one or more orders of magnitude stronger than the RF signal, this self-rectification and nearby interference introduce tremendous DC-offset noise after direct-conversion. Furthermore, the amount of DC-offset generated by the local oscillator radiation is difficult to predict since its magnitude changes with receiver location and orientation. Good circuit isolation techniques could reduce this effect to a certain extent, but it cannot be eliminated entirely. The DC offset problem is particularly acute in GSM systems. GSM uses Gaussian minimum shift keying (GMSK) modulation which has a peak at DC. The offsets generated, directly add to the spectral peak of the downconverted signal. Generally, these offsets are much larger than the RMS front-end noise, severely degrading the SNR of the receiver. One way of removing the offsets is to use AC coupling. This, however, would require impractically large capacitors in order not to remove the signal bearing spectrum around DC.
In one approach to eliminating the DC offset, an analog DC notch filter implemented as simple capacitive coupling can be used to remove most of the DC offset noise. This approach, however, can only be used when the information bearing signal is not at or near DC as is the case, for example, of simple two-tone signaling, i.e. FSK modulation whose spectrum comprises little energy at DC.
When a portion of the information signal is, however, at or near DC, this approach cannot be used. Many of the modulation schemes used in personal communications require low frequency information to achieve low bit error rates. Thus, simple two-tone signaling is not suitable for most RF communications applications since these applications require modulation schemes that are more spectrally efficient than FSK. In these other modulation schemes, the method of capacitively coupling the baseband signal before sampling fails because of the large signal-bearing spectrum near DC, and a notch filter at DC will remove significant portions of the signal.
Multiplicative impairments manifest as distortion of the I and Q phasor results when the components of the I and Q channels are either uniformly distorted or differentially distorted. To avoid this type of distortion requires that all the components in the I and Q channel paths have constant gain and phase characteristics across their dynamic range. To assure such constant gain and phase characteristics is difficult and costly.
Another source of multiplicative type noise is the sampling jitter of the sample and hold used at the input to the A/D converters which introduces differential phase distortion between the I and Q channels.
A block diagram illustrating the impairment model of the channel and the receiver front end circuit is shown in FIG. 2. The channel is modeled as a dispersive, fading channel 42. The phase/frequency offset manifests as multiplicative noise 44. The I/Q mismatch 46 results from nonsymmetrical low pass filtering. The impairment is modeled as multiplying 54 the imaginary signal Q by an attenuation mismatch g while the real signal is considered the unimpaired reference. The real and impaired imaginary signals are added via adder 58. The DC offset additive impairment is then added to the signal via adder 60.
A diagram illustrating the effects of the noise impairments including I/Q gain mismatch and DC offset on an 8-PSK constellation is shown in FIG. 3. The effects of the multiplicative and additive impairments are illustrated with respect to an 8-PSK signal. The ideal unimpaired 8-PSK constellation 70 is a circle centered at the origin in the I/Q plane. The multiplicative impairments (i.e. attenuation mismatch by the LPFs, sampling jitter, etc.) impart an elliptical shape to the constellation. DC offset impairments further impart a linear shift to the constellation making the center of the constellation a point other than the origin.
It is therefore desirable to have a mechanism for compensating for the multiplicative noise impairments, particularly I and Q gain mismatch, in a communications receiver that is efficient, relatively easy to implement either in hardware, software or a combination of the two and is not costly to implement. An I/Q gain mismatch correction mechanism would permit the use of protocols and modulation schemes that have significant portions of their spectrum around or near DC such as GMSK and 8-PSK in EDGE based GSM systems.